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MIC3230/1/2
Constant Current Boost Controller for Driving High Power LEDs Bringing the Power to LightTM
General Description
The MIC3230/1/2 are constant current boost switching * 6V to 45V input supply range controllers specifically designed to power one or more strings of high power LEDs. The MIC3230/1/2 have an * Capable of driving up to 70W input voltage range from 6V to 45V and are ideal for a * Ultra low EMI via dithering on the MIC3231 variety of solid state lighting applications. * Programmable LED drive current The MIC3230/1/2 utilizes an external power device which * Feedback voltage = 250mV 3% offers a cost conscious solution for high power LED * Programmable switching frequency (MIC3230/1) or applications. The powerful drive circuitry can deliver up to 400kHz fixed frequency operation (MIC3232) 70W to the LED system. Power consumption has been minimized through the implementation of a 250mV * PWM Dimming and separate enable shutdown feedback voltage reference providing an accuracy of 3%. * Frequency synchronization with other MIC3230s The MIC323x family is dimmable via a pulse width * Protection features: modulated (PWM) input signal and also features an enable Over Voltage Protection (OVP) pin for low power shutdown. Over temperature protection Multiple MIC3230 ICs can be synchronized to a common Under-voltage Lock-out (UVLO) operating frequency. The clocks of these synchronized * Packages: devices can be used together in order to help reduce noise and errors in a system. VIN 1 10 VDD An external resistor sets the adjustable switching EN 2 9 DRV frequency of the MIC3230/1. The switching frequency can PWMD 3 8 PGND COMP 4 7 OVP be between 100kHz and 1MHz. Setting the switching IADJ 5 6 IS frequency provides the mechanism by which a design can be optimized for efficiency (performance) and size of the MIC3232 MIC3230/1 MIC3230/1 external components (cost). The MIC323x family of LED MSOP-10 MLF-12 TSSOP-16 drivers also offer the following protection features: Over voltage protection (OVP), thermal shutdown and under* -40C to +125C junction temperature range voltage lock-out (UVLO). The MIC3231 offers a dither feature to assist in the reduction of EMI. This is particularly useful in sensitive Applications EMI applications, and provides for a reduction or * Street Lighting emissions by approximately 10dB. * Solid State Lighting The MIC3232 is a 400kHz fixed frequency device offered * General Illumination in a small MSOP-10 package. The MIC3230/1 are offered in both the EPAD TSSOP-16 package and the * Architectural Lighting (R) 3mm x 3mm MLF -12 package. * Constant Current Power Supplies Data sheets and support documentation can be found on Micrel's web site at: www.micrel.com. _________________________________________________________________________________________________
N/C 1 16 N/C
VIN EN 1 2 3 12 VDD 11 DRV
Features
VIN EN
2 3 4 5 6 7 8
15 VDD 14 DRV 13 PGND 12 OVP 11 IS
PWMD
10 PGND 9 8 7 OVP
PWMD COMP
COMP IADJ FS
4
5 6
IS
IADJ
EPAD
SYNC/NC
FS
10 SYNC/NC 9 N/C
AGND
EPAD
Bringing the Power to Light is a trademark of Micrel, Inc. MicroLead Frame and MLF are registered trademark of Amkor Technologies. Micrel Inc. * 2180 Fortune Drive * San Jose, CA 95131 * USA * tel +1 (408) 944-0800 * fax + 1 (408) 474-1000 * http://www.micrel.com
January 2009
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Micrel, Inc.
MIC3230/1/2
Typical Application
L 47H VIN CIN 4.7F/50v R2 100k ENABLE PWMD Synch to other MIC3230 VIN EN PWMD SYNC MIC3230/31 FS COMP RFS 16.5k CCOMP 10nF VDD C3 10F 10V Analog ground AGND IS IADJ PGND RSLC 51 VFB = 0.25V RADJ 1/4W Power ground RCS 1/2W DRV Q1 R9 4.33k LED X ILED Return OVP R8 100k COUT 4.7F 100V LED 1 D1 VOUT
EPAD
Figure 1. Typical Application of the MIC3230 LED Driver
Product Option Matrix
MIC3230 Input Voltage Synchronization Dither Frequency Range Package 6V to 45V Yes No Adj from 100kHz to 1MHz EPAD TSSOP-16 3mm x 3mm MLF(R)-12 MIC3231 6V to 45V No Yes Adj from 100kHz to 1MHz EPAD TSSOP-16 3mm x 3mm MLF(R)-12 MIC3232 6V to 45V No No Fixed Freq. = 400kHz MSOP-10
Ordering Information
Part Number MIC3230YTSE MIC3230YML MIC3231YTSE MIC3231YML MIC3232YMM Temperature Range -40 to +125C -40 to +125C -40 to +125C -40 to +125C -40 to +125C Package EPAD TSSOP-16 3mm x 3mm MLF -12L EPAD TSSOP-16 3mm x 3mm MLF -12L MSOP-10
(R) (R)
Lead Finish Pb-Free Pb-Free Pb-Free Pb-Free Pb-Free
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Pin Configuration
N/C 1 2 3 4 5 6 7 8 EPAD 16 N/C 15 VDD 14 DRV 13 PGND 12 OVP 11 IS 10 SYNC/NC 9 N/C
VIN 1 EN 2 PWMD 3 COM 4 P IADJ 5
10 VDD 9 DRV 8 PGND 7 OVP 6 IS
VIN EN PWMD COMP IADJ FS
1 2 3 4 5 6 EPAD
12 VDD 11 DRV
VIN EN PWMD COMP IADJ FS AGND
10 PGND 9 8 7 OVP IS SYNC/NC
MSOP-10 (MM) MIC3232
3mmx3mmMLF(R)-12L (ML) MIC3230, MIC3231 See Product Option Matrix for selection
TSSOP-16 (TSE) MIC3230, MIC3231 See Product Option Matrix for selection
Pin Description
Pin Number 3x3MLF -1 2 Pin Number TSSOP-16L 1 2 3 Pin Number MSOP-10L -1 2 NC VIN EN No Connect Input Voltage (power) 6V to 45V Enable Control (Input). Logic High (1.5V) enables the regulator. Logic Low (0.4V) shuts down the regulator. Connect a 100k resistor from EN to VIN. PWM input. High signal terminates the output power. Low Signal starts up the output power. Compensation (output): for external compensation Feedback (input) Frequency Select (input). Connected to a Resistor to determine the operating frequency Analog Ground No Connect Sync (output). Connect to another MIC3230 to synchronize multiple converters. Current Sense (input). Connected to external current sense resistor which in turn is connected to the source of the external FET as well as an external slope compensation resistor OVP divider connection (output). Connect the top of the divider string to the output. If the load is disconnected, the output voltage will rise until OVP reaches 1.25V and then will regulate around this point Power Ground Drive Output: connect to the gate of external FET (output) VDD Filter for internal power rail. Do not connect an external load to this pin. Connect 10F to GND. No Connect Connect to AGND Pin Name Pin Function
3 4 5 6 --7 8
4 5 6 7 8 9 10 11
3 4 5 ----6
PWMD COMP IADJ FS AGND NC SYNC IS
9
12
7
OVP
10 11 12 ---
13 14 15 16 --
8 9 10 ---
PGND DRV VDD NC EPAD
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Absolute Maximum Ratings(1)
Supply Voltage (VIN) .....................................................+48V Enable Pin Voltage........................................... -0.3V to +6V IADJ Voltage ..................................................................+6V Lead Temperature (soldering, #sec.)......................... 260C Storage Temperature (Ts)..........................-65C to +150C ESD Rating(3) ..................... MIC3230= 1500V HB, 100VMM .........................................MIC3232= 2kV HB, 100VMM .................................... MIC3231= 1500V HB, 150VMM
Operating Ratings(2)
Supply Voltage (VIN)......................................... +6V to +45V Junction Temperature (TJ)........................ -40C to +125C Junction Thermal Resistance MSOP-10 (JA) ..............................................130.5C/W EPAD TSSOP-16 (JA) ...................................36.5C/W 3mmx3mm MLF(R)-12L (JA).............................60.7C/W
Electrical Characteristics(4)
VIN = 12V; VEN = 3.6V; L = 47H; C = 4.7F; TJ = 25C, Bold values indicate -40C TJ +125C, unless noted. Symbol VIN UVLO IVIN ISD VIADJ IADJ Parameter Supply Voltage Range Under Voltage Lockout Quiescent Current Shutdown Current Feedback Voltage (at IADJ) Feedback Input Current Line Regulation Load Regulation DMAX VEN IEN VPWM fPWMD fSW Maximum Duty Cycle Enable Threshold Enable Pin Current PWMD Threshold PWMD Frequency Range Programmable Oscillator Frequency Fixed Frequency Option Low EMI (MIC3231) Current Limit Threshold Voltage ISENSE peak current out Over Voltage Protection Driver Impedance SINK SOURCE VFB > 275mV (to ensure device is not switching) VEN = 0V Room temperature (3%) -40C TJ +125C (5%) VFB = 250mV VIN = 12V to 24V VOUT to 2 x VOUT MIC3230 & MIC3232 MIC3231 Turn ON Turn OFF VEN = 3.3V REN = 100k Turn ON Turn OFF Note 5 (L = 47H; C = 4.7F) RFREQ = 82.5k RFREQ = 21k RFREQ = 8.25k (MIC3232YMM) Frequency dither shift from nominal RSENSE = 390 RSENSE = 390 1.203 0.315 1.5 0 360 360 109 400 950 400 12 0.45 250 1.24 2.4 2 1.277 3.5 0.585 90 88 1.5 1.15 1.1 17 0.75 0.7 0.4 30 242.5 237.5 Condition Min 6 3.5 4.9 3.2 30 250 250 1.2 2 2 257.5 262.5 3 Typ Max 45 5.5 10 Units V V mA A mV mV A % % % % V V A V V Hz kHz kHz kHz kHz % V A V
0.4 500 440 440
fSW FDITHER VSENS ISENSE VOVP
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Micrel, Inc. VDRH TJ Driver Voltage High Over-Temperature Threshold Shutdown Thermal Shutdown
Notes: 1. Exceeding the absolute maximum rating may damage the device. 2. The device is not guaranteed to function outside its operating rating. 3. Devices are ESD sensitive. Handling precautions recommended. Human body model, 1.5k in series with 100pF. 4. Specification for packaged product only. 5. Guaranteed by design
MIC3230/1/2 VIN = 12V 7 9 150 Hysteresis 5 11 V C C
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Typical Characteristics
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12.2 OUTPUT VOLTAGE (V) 12.15 12.1 12.05 12 11.95 11.9 11.85 11.8 0
Load Regulation
V 25
IN
= 3.6V
50 75 100 125 150 LOAD (mA)
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MIC3230/1/2 design flexibility in adjusting the current for a particular application need. The MIC3230/1/2 features a low impedance gate driver capable of switching large MOSFETs. This low impedance helps provide higher operating efficiency. The MIC323x family can control the brightness of the LEDs via its PWM dimming capability. Applying a PWM signal (up to 500Hz) to the PWMD pin allows for control of the brightness of the LED. Each member of the MIC323x family employs peak current mode control. Peak current mode control offers advantages over voltage mode control in the following manner. Current mode control can achieve a superior line transient performance compared to voltage mode control and through small signal analysis (not shown here), current mode control is easier to compensate than voltage mode control, thus allowing for a less complex control loop stability design. Figure 2 shows the functional block diagram.
Functional Description
A constant output current converter is the preferred method for driving LEDs. Small variations in current have a minimal effect on the light output, whereas small variations in voltage have a significant impact on light output. The MIC323x family of LED drivers are specifically designed to operate as constant current LED Drivers and the typical application schematic is shown in Figure 1. The MIC323x family are designed to operate as a boost controller, where the output voltage is greater than the input voltage. This configuration allows for the design of multiple LEDs in series to help maintain color and brightness. The MIC323x family can also be configured as a SEPIC controller, where the output voltage can be either above or below the input voltage. The MIC3230/1/2 have a very wide input voltage range, between 6V and 45V, to help accommodate for a diverse range of input voltage applications. In addition, the LED current can be programmed to a wide range of values through the use of an external resistor. This provides
VIN VDD OVP PWMD L1 VIN EN VOUT SYNC LED FS SYNC FS Clock Out VRAMP Out LDO VDD Internal VBIAS VOUT Control DRV Q1 C5
LED IADJ VC RADJ COMP VS 0.25V 10k
S Q R 0.45V Leading Edge Blanking Ai GND Ai = 1.4V ISLOPE COMP VRAMP IN
T 250a/T T
IS
RSLC RCS
ISLOPE COMP OUT
Figure 2. MIC3230 Functional Block Diagram
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MIC3230/1/2 in regulation the voltage at IADJ will equal 0.25V. Output Over Voltage Protection (OVP) The MIC323x provides an OVP circuitry in order to help protect the system from an overvoltage fault condition. This OVP point can be programmed through the use of external resistors (R8 and R9 in Figure 1). A reference value of 1.245V is used for the OVP. Equation 3 can be used to calculate the resistor value for R9 to set the OVP point. Eq. (3)
R9 = R8 (VOVP / 1.245) - 1
Power Topology
Constant Output Current Controller The MIC323x family are peak current mode boost controllers designed to drive high power LEDs. Unlike a standard constant output voltage controller, the MIC323x family has been designed to provide a constant output current. The MIC323x family is designed for a wide input voltage range, from 6V to 45V. In the boost configuration, the output can be set from VIN up to 100V. As a peak current mode controller, the MIC323x family provides the benefits of superior line transient response as well as an easier to design compensation. This family of LED drivers features a built-in soft-start circuitry in order to prevent start-up surges. Other protection features include: * Current Limit (ILIMIT) - Current sensing for over current and overload protection * Over Voltage Protection (OVP) - Output over voltage protection to prevent operation above a safe upper limit * Under Voltage Lockout (UVLO) - UVLO designed to prevent operation at very low input voltages Setting the LED Current The current through the LED string is set via the value chosen for the current sense resistor, RADJ. This value can be calculated using Equation 1: Eq. (1)
LED Dimming The MIC323x family of LED drivers can control the brightness of the LED string via the use of pulse width modulated (PWM) dimming. A PWM input signal of up to 500Hz can be applied to the PWM DIM pin (see Figure 1) to pulse the LED string ON and OFF. It is recommended to use PWM dimming signals above 120Hz to avoid any recognizable flicker by the human eye. PWM dimming is the preferred way to dim a LED in order to prevent color/wavelength shifting, as occurs with analog dimming. The output current level remains constant during each PWMD pulse. Oscillator and Switching Frequency Selection The MIC323x family features an internal oscillator that synchronizes all of the switching circuits internal to the IC. This frequency is adjustable on the MIC3230 and MIC3231 and fixed at 400kHz in the MIC3232. In the MIC3230/1, the switching frequency can be set by choosing the appropriate value for the resistor, R1 according to Equation 4: Eq. (4)
ILED =
0.25V R ADJ
Another important parameter to be aware of in the boost controller design, is the ripple current. The amount of ripple current through the LED string is equal to the output ripple voltage divided by the LED AC resistance (RLED - provided by the LED manufacturer) plus the current sense resistor (RADJ). The amount of allowable ripple through the LED string is dependent upon the application and is left to the designer's discretion. This equation is shown in Equation 2: Eq. (2) Where
ILED VOUTRIPPLE (RLED + R ADJ )
7526 RFS (k) = F (kHz ) SW
1.035
VOUTRIPPLE =
I LED x D x T COUT
Reference Voltage The voltage feedback loop of the MIC323x uses an internal reference voltage of 0.25V with an accuracy of 3%. The feedback voltage is the voltage drop across the current setting resistor (RADJ) as shown in Figure 1. When January 2009 9
SYNC (MIC3230 Only) Multiple MIC3230 ICs can be synchronized by connecting their SYNC pins together. When synchronized, the MIC3230 with the highest frequency (master) will override the other MIC3230s (slaves). The internal oscillator of the master IC will override the oscillator of the slave part(s) and all MIC3230 will be synchronized to the same master switching frequency. The SYNC pin is designed to be used only by other MIC3230s and is available on the MIC3230 only. If the SYNC pin is being unused, it is to be left floating (open). In the MIC3231, the SYNC pin is to be left floating (open).
M9999-011409-A
Micrel, Inc. Dithering (MIC3231 Only) The MIC3231 has a feature which dithers the switching frequency by 12%. The purpose of this dithering is to help achieve a spread spectrum of the conducted EMI noise. This can allow for an overall reduction in noise emission by approximately 10dB. Internal Gate Driver External FETs are driven by the MIC323x's internal low impedance gate drivers. These drivers are biased from the VDD and have a source resistance of 2 and a sink resistance of 3.5. VDD VDD is an internal linear regulator powered by VIN and VDD is the bias supply for the internal circuitry of the MIC323x. A 10F ceramic bypass capacitor is required at the VDD pin for proper operation. This pin is for filtering only and should not be utilized for operation. Current Limit The MIC323x family features a current limit protection feature to prevent any current runaway conditions. The current limit circuitry monitors current on a pulse by pulse basis. It limits the current through the inductor by sensing the voltage across RCS. When 0.45V is present at the IS pin, the pulse is truncated. The next pulse continues as normally until the IS pin reaches 0.45V and it is truncated once again. This will continue until the output load is decreased. Select RCS using Equation 5: Eq. (5)
MIC3230/1/2 Current Sense IS The IS pin monitors the rising slope of the inductor current (m1 in Figure 5) and also sources a ramp current (250A/T) that flows through RSLC that is used for slope compensation. This ramp of 250A per period, T, generates a ramped voltage across RSLC and is labeled VA in Figure 3. The signal at the IS pin is the sum of VCS + VA (as shown in Figure 3). The current sense circuitry and block diagram is displayed in Figure 4. The IS pin is also used as the current limit (see the previous section on Current Limit).
Figure 3. Slope compensation waveforms
RCS =
(V
OUTMAX
0.45 - VIN MIN x D
)
L x FSW
+ I LPK _ LIMIT
Soft Start The boost switching convertor features a soft start in order to power up in a controlled manner, thereby limiting the inrush current from the line supply. Without this soft start, the inrush current could be too high for the supply. To prevent this, a soft start delay can be set using the compensation capacitor (CCOMP in Figure 1). For switching to begin, the voltage on the compensation cap must reach about 0.7V. Switching starts with the minimum duty cycle and increases to the final duty cycle. As the duty cycle increases, VOUT will increase from VIN to it's final value. A 6A current source charges the compensation capacitor and the soft start time can be calculated in Equation 7: Eq. (7) TSOFTSTART
CCOMP x VCOMP_STEADY_STATE 6 A
Slope Compensation The MIC323x is a peak current mode controller and requires slope compensation. Slope compensation is required to maintain internal stability across all duty cycles and prevent any unstable oscillations. The MIC323x uses slope compensation that is set by an external resistor, RSLC. The ability to set the proper slope compensation through the use of a single external component results in design flexibility. This slope compensation resistor, RSLC, can be calculated using Equation 6: Eq. (6) RSLC
VCOMP_STEADY_STATE is usually between 0.7V to 3V, but can be as high as 5V. Eq. (8) VCOMP _ STEADY _ STATE = Ai x V APK + Vcs PK Where: V APK =
(
)
I RAMP x RSLC x D x T and T
(VOUT =
L x 250 A x FSW
MAX
- VIN MIN x RCS
)
VCSPK = IL _ PK x RCS
where VIN_MAX and VOUT_MAX can be selected to system specifications.
Ai = 1.4 V/V D = Duty cycle (0 to1) T = period A 10nF ceramic capacitor will make this system stable at all operating conditions.
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Micrel, Inc. Leading Edge Blanking Large transient spikes due to the reverse recovery of the diode may be present at the leading edge of the current sense signal. (Note: drive current can also cause such spikes) For this reason a switch is employed to blank the first 100ns of the current sense signal. See Figure 6.
I IN _ RMS = VOUT x I OUT eff x VIN
VIN
MIC3230/1/2
Eq. (10) Eq. (11)
IIN _ AVE =
(IIN _ PP )2 (IIN _ RMS ) - 12
2
2 Note: If IIN_PP is small then IIN_AVE nearly equals IIN_RMS
IIN _ PEAK = IIN _ AVE +
IIN _ PP
Eq. (9)
L1 Clock S Q R IL
250a/T
D1 DRV
PWM Comparator Ai
IS
VA +RSLC- VA = IRAMP x RSLP
VCS + RCS -
Current Limit 0.45V
VCS = IL x RCS
VC
0.45V IADJ
RCOMP = 10k COMP CCOMP
Figure 4. Current sense circuit (An explanation of the IS pin)
Clock T (1-D)T PWM
VC
DT IL_PK = IL_AVE + 1/2 IL_PP IL_AVE = IIN_AVE
m2 m1
IL
0
IL_PP
VC
IFET
0 VC
IL_AVE = IIN_AVE IFET_RMS
IDIODE
0
IOUT
Figure 5. Current Waveforms
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Figure 6. IS pin and VRCS (Ch1 = Switch Node, Ch2 = IS pin, Ref1 = VCS) Design Procedure for a LED Driver
Symbol VIN IIN
Parameter Input Input Voltage Input current Output
Min 8
Nom 12
Max 14 2
Units V A
LEDs VF VOUT ILED IPP PWMD OVP
Number of LEDs Forward voltage of LED Output voltage LED current Required I Ripple PWM Dimming Output over voltage protection System
5 3.2 16 0.33 0
6 3.5 21 0.35 40
7 4.0 28 0.37 100 V V A mA % V
30
FSW eff VDIODE
Switching frequency Efficiency Forward drop of schottky diode
500kHz 80 0.6 % V
Table 2. Design example parameters
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L 47H VIN CIN 4.7F/50v R2 100k ENABLE PWMD Synch to other MIC3230 VIN EN PWMD SYNC MIC3230/31 FS COMP RFS 16.5k VDD C3 10F 10V Analog ground AGND IS IADJ PGND RSLC 51 VFB = 0.25V RADJ 1/4W Power ground RCS 1/2W DRV Q1 OVP
D1 VOUT R8 100k COUT 4.7F 100V LED 1
LED X R9 4.33k ILED Return
CCOMP 10nF
EPAD
Figure 7. Design Example Schematic
Design Example In this example, we will be designing a boost LED driver operating off a 12V input. This design has been created to drive six LEDs at 350mA with a ripple of about 12%. We are designing for 80% efficiency at a switching frequency of 500kHz. Select RFS To operate at a switching frequency of 500kHz, the RFS resistor must be chosen using Equation 3.
= 16.6k 500 Use the closest standard value resistor of 16.5k.
Operating Duty Cycle The operating duty cycle can be calculated using Equation 12 provided below:
Eq. (12)
D=
(Vout - eff x Vin + Vdiode ) Vout + Vdiode
These can be calculated for the nominal (typical) operating conditions, but should also be understood for the minimum and maximum system conditions as listed below.
Dnom = Dmax = Dmin = (Vout nom - eff x Vinnom + Vschottky ) Vout nom + Vschottky (Voutmax - eff x Vinmin + Vschottky ) Voutmax + Vschottky (Vout min - eff x Vin max + Vschottky ) Vout min + Vschottky
(7526 )1.035 RFS (k ) =
Select RADJ Having chosen the LED drive current to be 350mA in this example, the current can be set by choosing the RADJ resistor from Equation 1:
Therefore DNOM =56% DMAX = 78% and DMIN = 33%
Inductor Selection First, it is necessary to calculate the RMS input current (nominal, min and max) for the system given the operating conditions listed in the design example table. This minimum value of the RMS input current is necessary to ensure proper operation. Using Equation 9, the following values have been calculated:
IIN _ RMS _ max = VOUT _ max x IOUT _ max eff x VIN _ min = 1.64 A _ rms
0.25V = 0.71 0.35 A The power dissipation in this resistor is: R ADJ =
P (R ADJ ) = I 2 * R ADJ = 87mW
Use a resistor rated at 1/4 watt or higher. Choose the closest value from a resistor manufacture.
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IIN _ RMS _ nom = IIN _ RMS _ min = VOUT _ nom x IOUT _ nom eff x VIN _ nom VOUT _ min x IOUT _ min eff x VIN _ max = 0.78 A _ rms = 0.48 A _ rms
MIC3230/1/2
Current Limit and Slope Compensation Having calculated the IL_pk above, We can set the current limit 20% above this maximum value:
I L _ pk Limit = 1.2 x 1.6 A = 1.9 A
Iout is the same as ILED Selecting the inductor current (peak-to-peak), IL_PP, to be between 20% to 50% of IIN_RMS_nom, in this case 40%, we obtain:
I in _ PP _ nom = 0.4I in _ rms _ nom = 0.4 * 0.78 = 0.31AP -P
The internal current limit comparator reference is set at 0.45V, therefore when VIS _ PIN = 0.45 , the IC enters current limit. Eq. (14)
0.45 = V APK + Vcs PK
(
)
(see the current waveforms in Figure 5). It can be difficult to find large inductor values with high saturation currents in a surface mount package. Due to this, the percentage of the ripple current may be limited by the available inductor. It is recommended to operate in the continuous conduction mode. The selection of L described here is for continuous conduction mode. Eq. (13)
V x D xT L = IN I in _ PP
12V x 0.56 x 2s = 43H 0.31A
Where V APK is the peak of the V A waveform and
Vcs PK is the peak of the Vcs waveform
Eq. (14a)
0.45 = I RAMP x RSLC x D + I L _ pk Limit x RCS
To calculate the value of the slope compensation resistance, RSLC, we can use Equation 5:
RSLC =
(VOUT
MAX
- VIN MIN x RCS
)
L x 250 A x FSW
Using the nominal values, we get:
L=
First we must calculate RCS, which is given below in Equation 15: 0.45 Eq. (15)
RCS =
(VOUTMAX - VINMIN ) x Dmax + I
L x FSW
L _ pk Limit
Select the next higher standard inductor value of 47H. Going back and calculating the actual ripple current gives: Eq. (13a)
Iin _ PP = VIN _ nom x Dnom x T L = 12v x 0.56 x 2us = 0.29APP 47uh
Therefore;
RCS =
(28v - 8v ) x (0.50) + 1.9 A = 179m
47 H x 500kHz
0.45
The average input current is different than the RMS input current because of the ripple current. If the ripple current is low, then the average input current nearly equals the RMS input current. In the case where the average input current is different than the RMS, Equation 10 shows the following: Eq. (13b)
IIN _ AVE _ max =
_ PP (IIN _ RMS _ max )2 - (IIN 12 ) 2
Using a standard value 150m resistor for RCS, we obtain the following for RSLC:
RSLC =
(28 - 8) x 150m = 511 47 H x 250 A x 500kHz
Use the next higher standard value if this not a standard value. In this example 511 is a standard value. Check: Because we must use a standard value for Rcs and RSLC; I L _ pk Limit may be set at a different level (if the calculated value isn't a standard value) and we must calculate the actual I L _ pk Limit value (remember
I L _ pk Limit is the same as I in _ pk Limit ).
I IN _ AVE _ max =
(1.64 )2 - (0.29)2 / 12 1.64 A
The Maximum Peak input current IL_PK can found using equation 11:
I L _ PK _ max = I IN _ AVE _ max + 0.5 x I L _ PP _ max = 1.78 A
The saturation current (ISAT) at the highest operating temperature of the inductor must be rated higher than this. The power dissipated in the inductor is: Eq. (13c) PINDUCTOR = Iin _ RMS _ max 2 x DCR
Rearranging Equation 14a to solve for I L _ pkLimit :
I in _ pkLimit = I in _ actual Limit =
(0.45 - I RAMP x RSLC x D ) RCS
(0.45 - 250ua x 511x 0.75) = 2.34 A .150
This is higher than the initial 1.2 x I L _ PK _ max = 1.9 A limit because we have to use standard values for RCS
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Micrel, Inc. and for RSLC. If I in _ actual Limit is too high than use a higher value for RCS. The calculated value of RCS for a 1.9A current limit was 179m. In this example, we have chosen a lower value which results in a higher current limit. If we use a higher standard value the current limit will have a lower value. The designer does not have the same choices for small valued resistors as with larger valued resistors. The choices differ from resistor manufacturers. If too large a current sense resistor is selected, the maximum output power may not be able to be achieved at low input line voltage levels. Make sure the inductor will not saturate at the actual current limit I in _ actual Limit . Perform a check at IIN=2.34Apk. VIS _ PIN = 250A x (0.78) x 511 + 2.34 A x 150m = 0.45V Maximum Power dissipated in RCS is; Eq. (17) Eq. (18)
PRCS = I RCS _ RMS 2 x RCS
IL _ PP 2 IRCS _ RMS _ max = IFET _ RMS _ max = D IIN _ AVE _ max 2 + 12
MIC3230/1/2 example of 6 LEDs, we obtain the following:
R LED _ total = 6 x 0.1 = 0.6
C out =
ILEDnom * Dnom * T = 4.1uF ILEDripple * (R adj + R LED _ total )
Use the next highest standard value, which is 4.7uF. There is a trade off between the output ripple and the rising edge of the PWMD pulse. This is because between PWM dimming pulses, the converter stops pulsing and COUT will start to discharge. The amount that COUT will discharge depends on the time between PWM Dimming pluses. At the next PWMD pulse COUT has to be charged up to the full output voltage VOUT before the desired LED current flows.
Input Capacitor The input current is shown in Figure 5. For superior performance, ceramic capacitors should be used because of their low equivalent series resistance (ESR). The input ripple current is equal to the ripple in the inductor plus the ripple voltage across the input capacitor, which is the ESR of CIN times the inductor ripple. The input capacitor will also bypass the EMI generated by the converter as well as any voltage spikes generated by the inductance of the input line. For a required VIN_RIPPLE: Eq. (21)
= 1.4F 8 x VIN _ RIPPLE x FSW 8 x 50mV x 500kHz This is the minimum value that should be used. The input capacitor should also be rated for the maximum RMS input current. To protect the IC from inductive spikes or any overshoot, a larger value of input capacitance may be required and it is recommended that ceramic capacitors be used. In this design example a value of 4.7F ceramic capacitor was selected. CIN = = I IN _ PP
0.26 2 = 1.44 A _ rms I RCS _ RMS = 0.781.64 2 + 12
PRCS = 1.25 2 x .15 = 0.31watt
Use a 1/2 Watt resistor for RCS.
Output Capacitor In this LED driver application, the ILED ripple current is a more important factor compared to that of the output ripple voltage (although the two are directly related). To find the COUT for a required ILED ripple use the following calculation:
(0.28 A)
For an output ripple ILEDripple = 20% of ILEDnom
ILED ripple = 0.2 x 0.35 = 70mA
Eq. (19)
C out =
ILED nom * D nom * T ILED ripple * (R adj + R LED _ total )
Find the equivalent ac resistance RLED _ ac from the datasheet of the LED. This is the inverse slope of the ILED vs. VF curve i.e.: Eq. (20)
MOSFET Selection In this design example, the FET has to hold off an output voltage maximum of 30V. It is recommended to use an 80% de-rating value on switching FETs, so a minimum of a 38V FET should be selected. In this design example, a 75V FET has been selected. The switching FET power losses are the sum of the conduction loss and the switching loss:
RLED _ ac =
VF ILED
Eq. (22)
PFET = PFET _ COND + PFET _ SWITCH
In this example use R LED _ ac = 0.1 for each LED. If the LEDs are connected in series, multiply R LED _ ac = 0.1 by the total number of LEDs. In this January 2009 15
The conduction loss of the FET is when the FET is turned on. The conduction power loss of the FET is found by the following equation: Eq. (23)
PFET _ COND = IFET _ RMS 2 x RDSON , where
M9999-011409-A
Micrel, Inc.
2 I L _ PP 2 = D I IN _ AVE + 12
MIC3230/1/2 From Equation 26: ttransition
Qg 68nC = = 34ns Igatedrv 2A
I FET _ RMS
I FET _ AVE _ max = 1.64 A
VOUT _ max = 28V
The switching losses occur during the switching transitions of the FET. The transition times, ttransition, are the times when the FET is turning off and on. There are two transition times per period, T. It is important not to confuse T (the period) with the transition time, ttransition. Eq. (24) Eq. (25)
PFET _ SWITCH _ max = IFET _ AVE _ max x VOUT _ max x ttransition _ max x FSW
From Equation 25:
PFET _ SWITCH _ max = 1.64 A x 28V x 34ns x 500 kHz = 0.78Watts
From Equation 22
PFET = 62mW + 0.78W = 0.84W
T=
1 Fsw
This about the limit for a part on a circuit board without having to use any additional heat sinks.
Rectifier Diode A Schottky Diode is best used here because of the lower forward voltage and the low reverse recovery time. The voltage stress on the diode is the max VOUT and therefore a diode with a higher rating than max VOUT should be used. An 80% de-rating is recommended here as well.
To find ttransition _ max : Eq. (26) ttransition _ max
Qg Igatedrv
where Qg is the total gate charge of the external MOSFET provided by the MOSFET manufacturer and the Qg should chosen at a VGS10V. This is not an exact value, but is more of an estimate of ttransition _ max . The FET manufacturers' provide a gate charge at a specified VGS voltage:
CIn _ FET = QG @VGS
Eq (28)
IL _ PP 2 Idiode _ RMS _ max = (1 - D ) IIN _ AVE _ max 2 + 12
Eq. (29)
Pdiode VSCHOTTKY x I diode _ RMS _ max Pdiode 0.81W
This is the FET's input capacitance. Select a FET with RDS(on) and QG such that the external power is below about 0.7W for a SO-8 or about 1W for a PowerPak (FET package). The Vishay Siliconix Si7148DP in a PowerPak SO-8 package is one good choice. The internal gate driver in the MIC3230/1/2 is 2A. From the Si7148DP data sheet: RDS(on)_25C=0.0145 Total gate Charge=68nC (typical) The R DS(on ) (temp ) is a function of temperature. As the temperature in the FET increases so does the RDS(on). To find RDS(on ) (temp ) use Equation 27, or simply double the R DS (on ) (25 o C ) for R DS (on ) (125 o C ) . Eq. (27) R DS(on ) (temp ) = R DS(on ) (25 o C ) x (1.007 (Temp -25 ) )
o
MIC3230 power losses
The power losses in the MIC3230are: Eq.(30)
PMIC 3230 = Qgate x Vgate x F + IQ x Vin
where Q gate is the total gate charge of the external MOSFET. Vgate is the gate drive voltage of the MIC3230. F is the switching frequency.
I Q is the
quiescent current of the MIC3230 found in the electrical characterization table. IQ = 3.2mA . VIN is the voltage at
the VIN pin of the MIC3230. From Eq.(30)
PMIC 3230 = 68nF x 12 x 500kHz + 3.2mA x 14 = 0.45W
The R DS (on ) (temp ) at 125C is:
R DSon (125 o C ) = 0.0145 x (1.007 (125
-25o )
) 30m
From Equation 23: PFET _ COND = 1.64 2 x 30m = 62mW
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Micrel, Inc.
OVP-Over voltage protection
MIC3230/1/2
Set OVP higher than the maximum output voltage by at least one volt. To find the resistor divider values for OVP use Equation 3 and set the OVP=30V and R8=100k:
R9 = 100k x 1.245 = 4.33k 30 - 1.245
2. Even though the RRC is very short (tens of nanoseconds) the peak currents are high (multiple amperes). The high RRC causes a voltage drop on the ground trace of the PCB and if the converter control IC is referenced to this voltage drop, the output regulation will suffer. 3. It is important to connect the IC's reference to the same point as the output capacitors to avoid the voltage drop caused by RRC. This is also called a star connection or single point grounding. 4. Feedback trace: The high impedance traces of the FB should be short.
PCB Layout
1. All typologies of DC-to-DC converters have a reverse recovery current (RRC) of the flyback or (freewheeling) diode. Even a Schottky diode, which is advertised as having zero RRC, it really is not zero. The RRC of the freewheeling diode in a boost converter is even greater than in the Buck converter. This is because the output voltage is higher than the input voltage and the diode has to charge up to -VOUT during each on-time pulse and then discharge to VF during the off-time.
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Micrel, Inc.
MIC3230/1/2
Package Information
10-Pin MSOP (MM)
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M9999-011409-A
Micrel, Inc.
MIC3230/1/2
(R) 12-Pin 3mm x 3mm MLF (ML)
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M9999-011409-A
Micrel, Inc.
MIC3230/1/2
16-Pin Exposed Pad TSSOP (TSE)
MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA
TEL +1 (408) 944-0800 FAX +1 (408) 474-1000 WEB http://www.micrel.com
The information furnished by Micrel in this data sheet is believed to be accurate and reliable. However, no responsibility is assumed by Micrel for its use. Micrel reserves the right to change circuitry and specifications at any time without notification to the customer. Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product can reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A Purchaser's use or sale of Micrel Products for use in life support appliances, devices or systems is a Purchaser's own risk and Purchaser agrees to fully indemnify Micrel for any damages resulting from such use or sale. (c) 2009 Micrel, Incorporated.
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M9999-011409-A

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